1. Field of the Invention
This invention relates to wireless communications. More specifically, this invention relates to systems for digital wireless communications that employ coherent detection.
2. Description of Related Art and General Background
1) Spread Spectrum and Code-division Multiple-access
Spread spectrum communication techniques are robust to noise, allow for the use of low transmission power, and have a low probability of intercept. For such reasons, much of the early development of spread spectrum technology was performed by military researchers. Recently, however, the advantages of this technology have led to its increasing use for consumer applications as well: most notably, in advanced digital cellular telephone systems.
Whereas most other communication techniques modulate a carrier signal with one or more data signals alone, spread spectrum techniques also modulate the carrier with a pseudorandom noise or xe2x80x98pseudonoisexe2x80x99 (PN) signal. In the frequency-hopping variant of spread spectrum systems, the value of the PN signal at a particular instant determines the frequency of the transmitted signal, and thus the spectrum of the signal is spread. In the direct sequence spread spectrum (DSSS) variant, the bit rate of the PN signal (called the xe2x80x98chip ratexe2x80x99) is chosen to be higher than the bit rate of the information signal, such that when the carrier is modulated by both signals, its spectrum is spread.
Communication systems that support multiple individual signals over a single channel must employ some technique to make the various signals distinguishable at the receiver. In time-division multiple-access (TDMA) systems, the individual signals are transmitted in nonoverlapping intervals such that they are orthogonal (and thus separable) in time space. In frequency-division multiple-access (FDMA) systems, the signals are bandlimited and transmitted in nonoverlapping subchannels such that they are orthogonal in frequency space. In code-division multiple-access (CDMA) systems, the signals are spread through modulation by orthogonal or uncorrelated code sequences such that they are orthogonal or nearly orthogonal in code space and may be transmitted across the same channel at the same time while remaining distinguishable from one another at the receiver. An exemplary CDMA system is described in U.S. Pat. No. 4,901,307, entitled xe2x80x9cSPREAD SPECTRUM MULTIPLE-ACCESS COMMUNICATION SYSTEM USING SATELLITE OR TERRESTRIAL REPEATERS,xe2x80x9d issued Feb. 13, 1990 and assigned to the assignee of the present invention, and the disclosure of which is hereby incorporated by reference.
In a CDMA DSSS system, then, each individual carrier signal is modulated by a data signal and a pseudonoise (PN) signal that is at least nearly orthogonal to the PN signals assigned to all other users, thus spreading the spectrum of the transmitted signal while rendering it distinguishable from the other users"" signals. Before spreading and modulation onto the carrier, the data signal typically undergoes various encoding and interleaving operations designed, for example, to increase data redundancy and allow error correction at the receiver. The data signals may also be encrypted to provide extra security against eavesdroppers. The generation of CDMA signals in a spread spectrum communications system is disclosed in U.S. Pat. No. 5,103,459, entitled xe2x80x9cSYSTEM AND METHOD FOR GENERATING SIGNAL WAVEFORMS IN A CDMA CELLULAR TELEPHONE SYSTEM,xe2x80x9d issued Apr. 7, 1992 and assigned to the assignee of the present invention, and the disclosure of which is hereby incorporated by reference.
2) Phase Modulation
In a DSSS telecommunications system, the baseband information signal is typically spread by the PN sequences to have a bandwidth of 1 MHz or more. In order to transmit the spread baseband signal over a radio channel, it is necessary to modulate it onto an RF carrier of the desired frequency.
Various methods of modulating digital baseband signals onto RF carriers exist. These methods typically operate by varying the amplitude, phase, and/or frequency of one or both of the in-phase (I) and quadrature (Q) components of the carrier according to the data symbol to be transmitted at any particular instant. DSSS systems commonly use a variant of either phase-shift keying (PSK), in which the phase states in the carrier components correspond to data symbols being transferred, or quadrature amplitude modulation (QAM), in which both the phases and the amplitudes of the carrier components are modulated.
In an exemplary system using binary PSK (BPSK) modulation, a transition of the carrier from a base phase state (defining a phase of zero) to a second phase state which is different by 180 degrees (i.e. a phase shift of xcfx80 radians away from zero) may be designated to indicate a transition from a data symbol 0 to a data symbol 1. The converse phase shift of xcfx80 radians back to zero would then be designated to indicate a transition from a data symbol 1 to a data symbol 0. Between these transitions, the phase of the carrier indicates whether a data symbol 0 is being transmitted (phase of zero) or a data symbol 1 instead (phase of xcfx80 radians). An improved ratio of data rate to bandwidth may be obtained by using quadrature PSK (QPSK) modulation, in which the data symbols are encoded into 180-degree shifts in both the I and Q components. These and other variants of PSK modulation are well known in the art.
Note that in PSK modulation, all phase states have significance only in relation to the base phase state. If this reference state is unknown, then only the points of phase state transition may be identified, and the actual identities of the symbols cannot be determined. In the BPSK system described above, for example, a phase shift of xcfx80 radians indicates either a transition from 0 to 1 or from 1 to 0. Unless one knows the relation between the base phase state and either the starting or the ending phase state, it is impossible to determine which transition was meant.
This phase ambiguity problem may be addressed in several different ways. One approach has been to avoid it by using a modulation which is suitable for noncoherent detection and does not require knowledge of the base phase state, such as differential PSK (DPSK). A more power-efficient method uses orthogonal signalling sets to encode the data symbols in a manner which is unambiguous regardless of the base phase state. The Hadamard-Walsh functions are one suitable signalling set, as discussed in Chapter 4 of CDMA: Principles of Spread Spectrum Communications by Andrew J. Viterbi, Addison Wesley Longman, Reading, Mass., 1995, which chapter is herein incorporated by reference. However, by providing the informational redundancy necessary to avoid the phase ambiguity problem, these methods may also reduce the achievable data throughput of the channel. An alternative approach has been to use a coherent detection scheme.
3) Coherent Detection and Phase Noise
In pilot-assisted coherent detection, the base phase state is derived from a pilot signal, a signal of known form which is transmitted along with the data signal to provide a phase and magnitude reference. One method of transmitting pilot and data channels over the same carrier is to cover the channels with different orthogonal codes (e.g., with different Walsh functions). At the receiver, the pilot channel may be used to establish carrier synchronization and enable coherent detection by, for example, using a phase-locked loop to keep the output of a local oscillator at a constant phase angle with respect to the received pilot.
Unfortunately, carrier synchronization is often complicated by the presence of phase noise. Phase noise may have two components, one being random and the other being more determinable. The random component is primarily due to Doppler effects caused by relative motion between the transmitter and receiver (or apparent motion between the two, as might be caused by a reflector). The maximum magnitude fdxe2x80x94max of such a Doppler shift is defined as
fdxe2x80x94max=fcxc3x97v/c,
where fc is the carrier frequency in Hz, v is the relative velocity in m/sec, and c is the speed of light. For carrier frequencies in the gigahertz range and relative velocities of a few hundred miles per hour, the Doppler component may be on the order of hundreds of Hertz.
Because the Doppler component changes rapidly and may exceed a few percent of the data rate, it is typically very difficult to track using a phase-locked loop. An alternative way to compensate for this random component is to use the known pilot signal to obtain an estimate of the channel""s effects. This estimate is usually in the form of a complex vector which represents the rotation in phase introduced by the channel and is used to compensate for the same rotation in the data samples.
Phase noise also arises as a frequency offset within the system architecture, caused primarily by a difference in frequency between the oscillators in the transmitter and the receiver. This difference may arise, for example, because of variances in manufacture or drift due to aging or temperature, and the effect of this offset is to introduce a phase rotation into the samples which remains relatively constant with respect to time. Section 10.1.1.3 of the IS-98A standard (TIA/EIA, July 1996) allows the frequency of the mobile station""s carrier to have an error of as much as 300 Hz once the oscillators have been phase-locked.
If the constant-rotating-component frequency offset is compensated separately, then a much better estimate of the random component of the phase noise may be obtained. One method of correcting the frequency offset is by using a digital frequency-locked loop (DFLL). The elements and principles of operation of digital frequency-locked loops are well known in the art and are described, for example, in xe2x80x9cConvergence and Output MSE of Digital Frequency-Locked Loop for Wireless Communicationsxe2x80x9d by the inventor Ling, Proceedings of the 1996 IEEE Vehicular Technology Conference, Atlanta, pp. 1215-1219, and xe2x80x9cAFC Tracking Algorithmsxe2x80x9d by Francis D. Natali, IEEE Transactions on Communications, vol. COM-32, no. Aug. 8, 1984, pp. 935-947, which documents are hereby incorporated by reference.
Frequency Offset Correction
FIG. 1 illustrates one method of frequency offset correction that uses a DFLL. An RF stage (not shown) supplies received analog data to conversion and correction block 110 for A/D conversion and frequency correction. As detailed below, the frequency correction operation may be performed either before or after A/D conversion. The digitized and corrected in-phase (I) and quadrature (Q) sample streams are despread by PN spreader 115 and then by data despreader 120 and pilot despreader 130 to obtain the symbols of the data and pilot channels, respectively.
Frequency discriminator 140 receives the despread pilot samples and generates an instantaneous frequency error {circumflex over (f)}. As illustrated in FIG. 2, the value {circumflex over (f)} is calculated as the imaginary portion of the complex product of the current pilot sample and the complex conjugate of the previous pilot sample. In loop filter 150, the instantaneous frequency error {circumflex over (f)} is scaled to control the convergence and bandwidth of the DFLL and then integrated to obtain a more accurate offset frequency estimate {tilde over (f)}. This estimate of offset frequency is input to block 110 and used to adjust the phase of the received data.
In block 110, frequency correction may be applied in the analog domain by inputting analog data at RF or IF and using a voltage-controlled oscillator (VCO) controlled by the DFLL to downconvert the signal to baseband before A/D conversion. While it is relatively easy to perform frequency correction on an analog signal, however, for some applications it is not practical. For example, the signal received by a CDMA base station will typically contain signals from many users, each component signal having a different frequency offset. For a TDMA base station, the signal received during each time slot will typically come from a different user and have a different frequency offset than the signal received in adjacent time slots. In such cases, it is preferable to perform the frequency correction in the digital domain. Moreover, better temperature stability and reliability of operation can be obtained in a smaller circuit area if the correction is performed digitally instead.
Frequency correction may be applied in the digital domain by inputting analog data to block 110 at baseband and applying complex rotations to the samples after A/D conversion. One disadvantage to performing the correction after digitization is that it becomes more intensive, requiring a complex rotation to be performed on every sample. For a typical chip rate of 1.2288 Mcps and a sampling rate of twice the chip rate, this method would require nearly 2.5 million complex rotations to be performed every second. The power and available area required to support such a processing rate renders digital frequency correction infeasible for many applications.
By performing frequency correction processing on the pilot channel after despreading, rather than on the received samples before despreading, the invention considerably reduces the computational effort required to compensate for a frequency offset in the data channel. In order to maintain coherence between the data and pilot channels, the invention also includes derotating the channel estimate and delaying the samples in the data channel before coherent detection is performed.